A typical isolated switching regulator, such as the forward converter of FIG. 1, has a single controllable switch on the primary side, while the controlling quantity, the desired regulated output voltage V1 of the main output is on the secondary side. This requires at least one crossing of the isolation barrier in the feedback circuitry. However, in practice, due to the additional requirements for floating drives, remote sensing etc., the feedback control and drive circuit contains typically five or more isolation transformer and/or optocouplers. Thus, it would be very beneficial to have the switching converter with a minimum number of isolation barrier crossings, and ideally only one in the main power transformer and none in the feedback control and drive circuitry. A number of feedback control methods with varying number of isolation crossings in the feedback path are discussed at length by Bob Mamano in Isolating the Control Loop, Unitrode Switching Power Supply Design Seminar Manual, 1991 edition. Thus, one objective of this invention is to create an isolated switching converter which completely eliminates the need for crossing the isolation barrier in the feedback control path and reduce the safety considerations just to the power transformer alone. Another objective is to realize this with simple and inexpensive control circuits.
A typical switching regulator, such as the forward converter of FIG. 1, is required to maintain the regulated output voltage despite the wide changes in the input voltage and wide load current changes. Of the two disturbances, input voltage variations have a much more significant effect on the duty ratio of the single controllable device, active switch Q on the primary side. For example, a 4 to 1 input voltage change (15 V to 60 V for example) may cause a change of duty ratio from 0.2 to 0.8 (or 4 times) in the forward converter due to its linear dc gain characteristic. On the other hand, a load change of 10 times (from 10% load to full load) causes only a fraction of the change of duty ratio, such as 0.05 or less as long as the converter remains in Continuous Inductor Current Mode (CICM). The small change in duty ratio is to just make a small adjustment in output voltage due to the voltage drop in resistive parasitic losses of the converter. Thus, for a 90% efficient converter, the duty ratio may experience only a 0.05 change, or an order of magnitude smaller change than for input voltage variations. Therefore, the separation of the two disturbances is one objective of the present invention, which also leads to other advantages explained below. The lack of such a separation leads to relatively poor transient response with large overshoot and oscillatory ringing of the output voltage due to the sudden step changes of the input voltage. Due to the limited bandwidth of the closed-loop regulator and wide range of duty ratio changes required, the output voltage can not be instantaneously updated to new steady-state voltage, but instead undergoes the oscillatory transients. This is clearly undesirable performance for most loads and should be either eliminated or minimized.
A dual output extension of the forward converter in FIG. 1 illustrates another drawback associated with the prior-art switching converters. When the load on the main output is so low (light load condition), such that the output inductor peak to peak ripple current is larger than twice the dc load current, the converter enters in a so-called Discontinuous Inductor Current Mode (DICM) of operation, in which the voltage gain is not only function of duty ratio, but also depends on the dc load current, inductance values, and switching frequency as described in Slobodan Cuk and R. D. Middlebrook, "A General Unified Approach to Modeling Switching DC-to-DC Converters in Discontinuous Conduction Mode", Proceedings of IEEE PESC Conference, June 1977. Although, the main output would be still regulated at its prescribed value, the duty ratio would change to accommodate this mode of operation. As a consequence the secondary output voltage would have decreased substantially from its nominal value. Similarly, when the second output enters the DICM mode, its voltage would increase. As a result, the nominal 12 V output could vary anywhere between 6 V and 18 V, which is unacceptable in practice. The brute force solution is to pre-load each output with some resistance, which is wasteful. Another solution shown in FIG. 1 is to add another buck switching regulator. Although more efficient, this adds to the size, weight and cost of the power supply and is generally not practiced except in extreme circumstances. The input power is still processed twice for the second output (cascade connection of forward and buck converters) still resulting in sizable efficiency degradation. For example, with 90% efficiency of each stage, the overall efficiency would be only 81%. Thus, another objective of this invention is to achieve the full regulation of all output voltages in a single power conversion stage with multiple outputs, but still preserve full regulation of all outputs from full load through light load to no load conditions as well as have independent protection of each output.
Output voltage regulation against variation of two quantities, that is input voltage variation and output load variation, is typically achieved using single feedback loop such as shown in FIG. 1. As a consequence, the response to the input voltage variations, especially over wide input range voltage, results in a transient in the output voltage with sizable overshoot and oscillations. One attempt to improve such response is to add feedforward compensation to the buck converter by modulating the slope of the otherwise fixed up-going ramp reference signal with the input voltage as shown in FIG. 2a. Thus, the range of duty cycle change is reduced and transient response to input voltage change improved. Due to the linearity of the buck converter dc gain characteristic and its simple two pole, minimum phase frequency response, this feed-forward control results in virtual elimination of the transient due to input voltage change, but only in the Continuous Inductor Current Mode (CICM) as seen in FIG. 2b. As soon as the converter enters DICM mode at light load and no load, the feed-forward control is ineffective and results in undesirable output voltage transients as shown in FIG. 2c. These feedforward advantages in CICM mode of operation are not limited to the buck converter. Despite, the non-linear dc gain characteristics of the boost and flyback converters, the similar advantages of improved input voltage transient response can be obtained by use of an appropriate optimum feedforward strategy, such as described in Lloyd Dixon, Jr. "Pulse Width Modulator Control Methods With Complementary Optimization", Power Conversion International Magazine, January 1982. However, as pointed by Dixon, once again, all advantages are lost at light load, when the converter enters DICM mode. In addition, as pointed in Dixon, the very unfavorable frequency response of the control to output transfer function of the flyback converter, having a non-minimum phase response and right half-plane zero, is not eliminated by this feedforward control, and is therefore plaguing this approach even in the CICM mode of operation. This is clearly the consequence of the fact that feedforward is always used in addition to and combined with the regular output voltage feedback control. Thus, regulation against the input voltage changes is even in this optimum feedforward control achieved via closed-loop output voltage feedback control with all limitations it imposes. Thus, one motivation of the present invention is to find a suitable switching converter configuration to regulate against input voltage changes in an open-loop fashion, which would entirely by-pass potential stability and oscillation problems as well as provide for an instantaneous and direct adjustment of the steady-state duty ratio in response to the sudden input voltage changes. The feedforward control of FIG. 2 still operates with two variations, that is input voltage variations and output load current variations still coupled through a single feedback loop and complex dynamics of the converter. Thus, another objective is to create a switching converter in which the input voltage variations are decoupled from the load current variations so that the ideal transient response of FIG. 2b for input voltage variations would be obtained for all load conditions from full load to no load, that is regardless of the conduction mode of operation together with the improved frequency response for both step input voltage and step load current changes.
Another deficiency of the switching converters is in high order dynamics and consequent complex and undesirable frequency response. For example, the switching converter disclosed in U.S. Pat. No. 4,184,197 and shown in FIG. 3 in its isolated configuration, exhibits at least a fourth order, non-minimum phase response shown in FIG. 4 due to the presence of the right half-plane (RHP) zero's in its loop-gain characteristic. This results in the 540% phase shift right after the first set of poles and zero's and results in difficulty in closing feedback loop without either damping or additional feedback control loop such as current-mode programming in which input switch current is used as a second feedback variable. Furthermore, the pole's and zero's are highly dependent on the steady-state duty ratio D, which varies over wide range for wide input voltage changes. Consequently, the feedback loop must be closed at lower frequency in order to insure operation under worst case condition and thus resulting in a sub-optimal bandwidth: typically 1 kHz for a 150 kHz switching frequency. Therefore, it would be very desirable to have a switching converter which exhibits an effective second order minimum phase response (no RHP zero's) loop-gain frequency response with a loop-gain bandwidth approaching the theoretical limit of one half the switching frequency, that is, 60 kHz bandwidth for 150 kHz switching frequency. This would then result in a small voltage overshoot with a fast settling time for step-load current changes even with a relatively small value of the output filtering capacitance. At present, a large output capacitor is used in order to reduce large output voltage overshoot due to step-load current change. Yet another objective is to provide the switching converter with the widest possible bandwidth for the load current regulation.
Several approaches have been proposed in the past which provided no-load to full-load regulation on all outputs in a single power conversion stage. The first approach described in A. Dauhajre and R. D. Middlebrook, "A Simple PWM-FM Control for Independently Regulated Dual Output Converter", Proc. Tenth International Solid State Power Electronics Conference (Powercon 10), March 1993, was based on a two-output flyback converter with one output operated in DICM mode, hence sensitive to switching frequency. The full regulation of two outputs was provided by controlling two quantities, duty ratio and the switching frequency of the single active device on the primary side. This method was clearly limited to the two outputs, required isolation in the feedback control circuit and operated at variable switching frequency, which is undesirable from Electromagnetic Interference (EMI) noise standpoint.
Another approach which achieves full regulation of all outputs but with a constant switching frequency is a Three-Switched Network converter shown in FIG. 5 and proposed by R. Mahadevan, S. El-Hamamsy, W. M. Polivka and S. Cuk in "A Converter With Three Switched Networks Improves Regulation, Dynamics, and Control", Proc. Tenth International Solid-State Power Electronics Conference (Powercon 10), March 1983. In this approach, the additional active switches Q.sub.1,D.sub.1 and Q.sub.2, D are added in each of the secondary circuits of a dual output isolated Three-Switched Network converter. For simplicity and noise reasons, all switches are synchronized and turned-ON at the same instant. If all active switches are also turned-OFF at the same instant, leading to identical duty ratios of all active switches, the circuit operation clearly reduces to that of the switching Cuk converter of FIG. 3. However, when the secondary side active switches are turned-OFF before the primary side active switch is turned-OFF, the resulting different duty ratios provide a means for an independent control of each of the outputs. Each of the two output voltages can be independently and fully controlled by separate PWM control signals q.sub.1 and q.sub.2.
In FIG. 5a, the arrows on the two drive waveforms q.sub.1 and q.sub.2 indicate that these edges are controllable. Note the wide range of change of both duty ratios d.sub.1 and d.sub.2. However, in the control strategy proposed in, R. Mahadevan, S. El-Hamamsy, W. M. Polivka and S. Cuk, "A Converter with Three Switched Networks Improves Regulation, Dynamics and Control," Proc. Powercom 10, 1983, the duty ratio d of the primary side transistor Q was constant, and its ability to vary was not utilized. Hence, both input voltage and load current variations are compensated by controlling only the duty ratios of active switches Q.sub.1 and Q.sub.2 on the secondary side of the transformer. Thus, the same dynamic response deficiencies and sub-optimal frequency response to either input voltage transient or step-load current transient remains. This is clearly the consequence of the complex fourth order frequency response which is further complicated by the presence of the undesirable right half-plane (RHP) zeros leading to the non-minimum phase response and potential stability problems. Moreover, wide input voltage range leads to large variation in the duty ratio and operating point and consequent heavily compromised bandwidth to insure stability under all operating conditions. Although not exposed to Mahadevan et al., the key advantage of the switching converter of FIG. 5 is that it obviates the need for isolation in the feedback control circuitry which was neither realized nor utilized in Mahadevan et al.